The paper considers devices designed to simulate a timefrequency structure of a radar signal re ected from a target or underlying surface. Proposed simulator can be used to simulate false radar targets including those located closer than the minimal delay the devices may implement in conventional simulators. An e ective way is suggested to reduce the simulated range. The method works by real time application of an additional frequency shift. It may be useful to simulate the combat operation of a radar system, as well as to simulate echo signals of radio altimeters when probing by signals with various types of linear frequency modulation. The dynamic modi cation of simulating parameters is carried out in accordance with parameters of the input emitted signal.
A device for simulating a false radar target is intended to simulate the timefrequency structure of a radar signal re ected from an underlying surface or from one or more targets located in a xed direction. It can be used to simulate the following echoes from false target including those located closer to the carrier, and from underlying surfaces for radio altimeters ( ight altitude meters).
Simulators may be useful to verify and even try to deceive the combat
operation of radar systems in di erent circumstances [
Here, radar systems, which operate with linear frequency modulated
continuous waves (FMCW) are discussed, because they are widely used in manned and
unmanned aircrafts and helicopters especially at low distances and altitudes.
Their signals are convenient to e ective processing with low level of emitted
power [
In order to qualitatively simulate the frequency-time structure of the re ected radar signals, we can use methods to form the re ected signal as a sum of signals re ected by number of di erent small at facets (areas) of the surface; such facets are su ciently smaller in comparison with the irradiated area. Facets may be equivalent to bright points in some target and surface models. Such approach is applicable and correct if their re ection coe cients are related to their physical areas, orientations, and local surface \brightnesses", i.e., the radar cross section corresponds to the carrier wavelengths.
A typical device for simulating radar targets with high resolution [
In practical application of the described methods and devices to simulating radar portraits with variable modulation parameters, the signi cant problem arises in simulating targets with ranges shorter than the distance to the carrier (or vehicle) that needs to be protected (hided) from the operating radar.
The same di culties appear when we use signal simulators to study the characteristics of radio altimeters in seminatural modeling of work in a laboratory. It is impossible to simulate a signal with a signal propagation delay smaller than the intrinsic delay caused by transition time in processing channels and waveguides.
Now, the minimum delay is 40 nsec even in the best known schemes of digital signal memory (DSM), which corresponds to the range of 6 m. With the use of ampli ers, lters, attenuators, and connecting cables in real simulators, the corresponding minimum simulated range is 14{60 m and more. It limits both the ability to hide the real position of the carrier protected from the high-resolution radar and the ability to simulate low altitudes when testing radio altimeters.
It is known that both the Doppler shift and the time delay of the signal, which
has a linear frequency modulation (FM) or an asymmetric sawtooth modulation,
can be simulated by a corresponding o set of its carrier frequency [
Therefore, to reduce the minimum simulated height and compensate its own delay in every hardware implementation, it is possible to use a certain frequency shift. For example, an FMCW altimeter with an up-chirp sawtooth modulation will register an equivalent low altitude if during the signal generation the additional frequency shift f is applied to the direction, which is approaching the transmitted ftx(t) and received frx(t) signals in the analysis window (measuring part of the signal inside each repetition interval). See Fig. 2 where the frx(t) graph is upshifted in the direction to be close to the ftx(t) graph.
Frequency of the heterodyning signal (so-called beat signal) at the output of the mixer in FMCW device is the di erence between the frequencies of the following two signals: f (t) = jftx(t) frx(t)j ' const = F b (1) where F b is the value of the beat frequency of the signal at the altimeter mixers output inside the analysis window excluding the transition windows (the beginning-and-ending periodical areas of modulation frequency jumps in ftx(t) and frx(t) graphs).
It is seen (Fig. 2) that the positive frequency shift f for the signal delayed by some minimum delay min will lead to a decrease in the average beat frequency F b and, as a result, to a decrease in the measured altitude or distance.
The rst disadvantage of this method is the dependence of the required sign of the frequency shift f on the direction (type) of the linear ramp fashion slope in up-chirp or down-chirp of the FMCW radar.
The second disadvantage is since the triangular wave modulation in the
FMCW, which is the most widely used scheme [
If the measurement unit of such radar operates on the leading edge of the
spectrum, then the problem of reducing the measured range could be solved.
But, let in the example of a typical FMCW device a radar system uses a range
measurement unit for the beat frequency corresponding to the \center of gravity"
of the spectrum averaged over the whole repetition interval [
The simple replacement of \+ f " by \{ f " will only lead to a change in the sequence of half-cycles with F bmin and F bmax, see the frx2(t) in Fig. 3. 3
For mentioned cases, it is suggested to generate a re ected signal as the sum of
two frequency-shifted copies of the emitted signal [
The frequency o set is performed of the same value f , but with opposite signs: \+ f " and \{ f ". Here, f is the parameter selected approximately equal to or larger than the selective locking/tracking lter in the radar receiver.
The simulator comprises a mixer and a delay forming unit. The fref is the frequency formed at the output of the mixer. With its help, the value of the rate of change of the modulation frequency (Vf = 2W=TM , where W is the modulation bandwidth and TM is the repetition interval, Fig. 3) can be evaluated in the delay forming device.
Further, a delay value for the adjustable delay line is calculated by using also the desired o set of the delay \ " coming from the external device = + f =Vf min = + f ref fref min; (2) where min is the own (internal) delay in the simulator circuits; D is the required o set delay with decreasing the simulated range or compensating for its own delay (when the value has the minus sign); ref is the value of the signal's taken delay from any convenient output of the multiple-tapped delay line (see Fig. 4).
So, the original signal in the adjustable delay line is delayed by the delay . In this case, the delay value determines the displacement of the entire portrait of the simulated target in the range to a smaller one (for < f =Vf ) or higher (for > f =Vf ). For example, when =0, the distance to the simulated target will be reduced by
Rmax = c 2min = c 2 Vf f ; where c is the speed of light.
The value of the parameter f is chosen approximately equal to or greater than the bandwidth of the selective capture and tracking lter in the radar (altitude in the altimeter). However, in compliance with the condition for the correct processing the received signal in the radar, 0 TM and, therefore, f TM Vf .
Thus by [
It is possible to modify the structure in order to improve the performance and
simplify the hardware implementation of the radar portraits simulator [
An example of decreasing the simulated range for the triangular FMCW is shown in Fig. 5. Here, we use the periodically changing the plus sign or the minus sign for a given frequency shift value f . So, the entire signal of the previously generated portrait of the simulated target takes the o set in its distance.
Periodic change of the sign is carried out on regular intervals, and we denote
this value t. Then, with continuous emission, the output signal will contain
t length segments of the emitted signal with phase jumps at the instant of
the sign change f . In the frequency domain, this will result in harmonics
corresponding to the sum, frequency di erences of the \useful" signal, and to the
sign changing frequency Fsign multiplied by an integer [
Let we select the Fsign (frequency of changing the sign) several times lower than the average carrier frequency f0, but above the band of \useful" signal modulation frequencies (for example, at t TM ) and take into account the actual presence of limiting frequency lters in all radar receivers. Under this the resulting signal in the working region will be equivalent by the spectral composition to the signal formed by the conventional summation of the signals, i.e., as in Fig. 4.
The principle of replacement of signal summation by signal \commutation"
was proposed and implemented in target simulators when probing with su
ciently long radiated signals [
Therefore, as a result of replacing the adder with the switchable modulator as in Fig. 4, we get a bifurcation of the spectra at each instant and in the rst and second half-periods. Two pairs of spectral envelopes (for the point target those are only harmonics F bmin and F bmax) in the general spectrum of S(f ) will be spaced along the frequency axis by 2 f , see Fig. 5. Thus, for the case of a FMCW radar with an arbitrary chirp slope, f is greater than the bandwidth of the bandpass (lowpass) beat signal lter. So, the high-frequency harmonics will be suppressed or discarded, because the \second simulated range" are far from the simulated target (surface). Moreover, and the real measured range value will correspond to the \center of gravity" of the low-frequency envelope of the spectrum with a smaller value of the \ rst simulated range" (altitude).
In Figure 5, the harmonic f min(t) are also shown for comparison. It corresponds to a signal with real minimum delay min with none of frequency shift. Its position in both half-periods of the modulation period is constant. So, it hits (Fig. 5) the tail part of the low-frequency envelope of the spectrum of the beat signal, i.e., to the right of the harmonic F bmin obtained by the proposed design of the simulator.
Thus, we simplify the hardware implementation of the simulator in Fig. 4 in the following way. Instead of the second adder and two frequency shifters, the sign switcher, one frequency shifter, and, also, additional adjustable ampli er are introduced. The device is shown in Fig. 6.
The simulator in Fig. 6 receives an emitted pulse from the radar, for which a radar portrait is created, by the receiving antenna through. Further there are the ampli er, multi-tapped delay line, set of modulators, adder, adjustable delay line, adjustable ampli er, and frequency shifter to the output of the simulator. The multi-tapped delay line provides simulation of bright points of the target(s) with individual delays. Individual amplitude and phase modulations are performed and they use the appropriate coe cients generated by the external device.
The output signal is obtained by shifting the frequency of the signal of the generated radar portrait of the target, and the frequency shift is performed alternately by the \+ f " and \{ f " units.
To implement this, the sign switcher is used. It inverts the frequency o set
value through time-equal intervals t, i.e., frequency of signal is changed at the
output of the sign switcher Fsign = 1=(2 t). Let the value of Fsign be selected
several times lower than the average carrier frequency f0 and outside the band
of the \useful" modulation frequencies of the radar signal; and let us take into
account the actual presence of limiting signal lters in all radar receivers. So,
the resulting signal in the working (usually low-frequency) region by the spectral
composition will be equivalent to the signal formed by the usual summation of
signals [
In the laboratory, as well as when working onboard with a low level of interference in the signal from the receiving antenna, it is possible to determine not only the absolute value of the rate of the linear frequency change but also the sign of that rate. This makes it possible to improve the quality characteristics by performing a frequency shift only in the desired direction. This means that harmonics corresponding to the second target will not be present in the output signal of such simulator.
To achieve this result, the simulator is supplemented with a chirps sign detector, whose input is fed from the ampli er (Fig. 7). The signal from the output of the chirps sign detector comes to an additional second input of the sign switcher of the frequency shift value; for example, this may be the value \ 1", where \+1" must correspond to \+ f " at the output of the sign switcher, and \{1", must match \{ f " at the output of the sign switcher.
When working on board of an aircraft, the level of interference and the level of the useful signal can vary within wide limits. This can lead to an incorrect work of the chirps sign detector and of the sign switch controlling the frequency shifting device. Therefore, it is necessary to complete the measurement of the parameters of the chirp signal by estimating the power characteristics of the signal from the receiving antenna in order to adjust the parameters of the simulator. To provide this, the simulator is supplemented with a Synchonizer-Detector (see Fig. 7).
The Synchonizer-Detector evaluates characteristics of the signal converted into the mixer and sets the corresponding mode of operation of the simulator. For example, it is provided the following way. If the signal level is insu cient, the simulator mode is selected with periodic change of the modulation frequency shift sign with equal intervals t independently of the output signal of the linear FM detector, and a gain value is given to the adjustable ampli er, for example, in the range A = 2:::3.
If the signal level is su cient, the chirps sign detector must correctly monitor the sign of the frequency change of the chirp signal. Therefore, such a mode of operation of the simulator is established, in which the modulation frequency shift is determined by the output signal of the chirps sign detector; here the gain value is given to the adjustable ampli er, for example, in the range A = 1:::1; 5.
The control of the gain value A allows one to improve the energy characteristics depending on the mode of operation of the simulator by reducing the level of the emitted signal of the target simulation by a factor of 2 with a correct determination of the sign of the change in the frequency of the chirp of the input signal. If there is no input signal with given/expected parameters or there is an intermittent or pulse radar signal, the gain value A = 0 can be output to the adjustable ampli er; this further improves the power characteristics.
To implement a delay line simulator device, modulators and adders can be analog or digital. To improve the quality of simulation, the signal generation is better to perform digitally on digital delay lines and modulators.
In [
In accordance with [
To exclude the output signal of the transmitting antenna from entering to the
input of the receiving antenna, it is possible to use a circulator, strobe operation,
and / or spatial diversity of the antennas [
Peculiarity of the described solution for simulator constructing is that regardless of the magnitude, direction, and combination of the signs of the rate of sawtooth frequency change, two identical false radar targets are simulated. The rst one is the main target. It can be simulated at a range shorter than the range of the radar carrier. The second target is assigned in range by 2 Rmax, and with the appropriate choice of the value of f , this will not interfere with the correct tracking for the main false radar target.
Acknowledgments. This work has been supported by the grant of the Ministry of education and science of the Russian Federation (project No8.2538.2017/4.6).